90-degree lumped and distributed Doherty impedance inverter

ABSTRACT

Apparatus and methods for a modified Doherty amplifier operating at gigahertz frequencies are described. The combining of signals from a main amplifier and a peaking amplifier occur prior to impedance matching of the amplifier&#39;s output to a load. An output impedance-matching element can be relied upon. In one example, the output impedance-matching element can include an output strip line, a shunt capacitor connected between the output strip line and ground, an output capacitor connected between the output strip line and an output bonding pad, and an inductive strip line connected between the output bonding pad and ground.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. Non-Provisional patentapplication Ser. No. 16/530,293 (“the '293 Application”), titled“90-Degree Lumped and Distributed Doherty Impedance Inverter,” filedAug. 2, 2019. The '293 Application is a Continuation-In-Part of andclaims priority to International Application No. PCT/IB2017/000727,titled “90-Degree Lumped and Distributed Doherty Impedance Inverter,”filed on Feb. 2, 2017. The '293 Application is also aContinuation-In-Part of and claims priority to International ApplicationNo. PCT/IB2017/000717, titled “Methods for Combining Doherty AmplifierSignals with 90-Degree Lumped and Distributed Impedance Inverters,”filed on Feb. 2, 2017. The '293 Application is also aContinuation-In-Part of and claims priority to International ApplicationNo. PCT/IB2017/000722, titled “90-Degree Lumped and Distributed DohertyImpedance Inverter,” filed on Feb. 2, 2017. Each of the foregoingapplications is hereby incorporated herein by reference in its entirety.

BACKGROUND Technical Field

The technology relates to high-speed, high-power, broad-bandwidth,integrated amplifiers which may be constructed from gallium nitridetransistors or transistors formed from other semiconductor materials.

Discussion of the Related Art

Gallium nitride semiconductor material has received appreciableattention in recent years because of its desirable electronic andelectro-optical properties. GaN has a wide, direct bandgap of about 3.4eV that corresponds to the blue wavelength region of the visiblespectrum. Light-emitting diodes (LEDs) and laser diodes (LDs) based onGaN and its alloys have been developed and are commercially available.These devices can emit visible light ranging from the violet to redregions of the visible spectrum.

Because of its wide bandgap, GaN is more resistant to avalanchebreakdown and can maintain electrical performance at higher temperaturesthan other semiconductors, such as silicon. GaN also has a highercarrier saturation velocity compared to silicon. Additionally, GaN has aWurtzite crystal structure, is a very stable and hard material, has ahigh thermal conductivity, and has a much higher melting point thanother conventional semiconductors such as silicon, germanium, andgallium arsenide. Accordingly, GaN is useful for high-speed,high-voltage, and high-power applications. For example, gallium-nitridematerials are useful in semiconductor amplifiers for radio-frequency(RF) communications, radar, RF energy, and microwave applications.

Applications supporting mobile communications and wireless internetaccess under current and proposed communication standards, such asWiMax, 4G, and 5G, can place austere performance demands on high-speedor RF amplifiers constructed from semiconductor transistors. Theamplifiers may need to meet performance specifications related to outputpower, signal linearity, signal gain, bandwidth, and efficiency.

SUMMARY

Methods and structures for improving the performance of high-speed,high-power, broad-band, integrated amplifiers are described. Thestructures and methods relate to circuitry for combining amplifiedsignals and impedance matching at the output of a modified Dohertyamplifier. Rearranging the order of signal combining and impedancematching compared to conventional Doherty amplifiers and using animpedance inverter that comprises an integrated distributed inductiveelement in the form of a microstrip line can appreciably improveamplifier bandwidth and allow scalability of signal amplification tohigher powers.

Some embodiments relate to a Doherty amplifier comprising an RF input, amain amplifier connected to the RF input, a peaking amplifier connectedto the RF input, a combining node at which an output from the mainamplifier combines with an output from the peaking amplifier, and animpedance inverter comprising an integrated distributed inductorconnected to an output of the main amplifier and to the combining node.

In some aspects, an input to the impedance inverter comprises one ormore bond wires connected to the output of the main amplifier. Thecombining node may be located at a drain bonding pad of the peakingamplifier. In some implementations, there are no impedance-matchingelements connected between the main amplifier and the impedance inverterto match an output impedance of the main amplifier to 50 ohms. Theimpedance inverter may rotate a phase of a first signal amplified by themain amplifier by no more than 95 degrees with respect to a phase of asecond signal amplified by the peaking amplifier.

According to some implementations, the distributed inductor comprises atleast one conductive strip line having a width and a length integratedon a substrate. The impedance inverter may further comprise lumpedinductive and capacitive elements. In some implementations, the stripline may be mainly inductive. In some cases, the strip line may bedivided in two and connected by a capacitor arranged in series. In someaspects, the impedance inverter may further comprise bond wires orconductive interconnects connected between the conductive strip line andoutputs of the main amplifier and the peaking amplifier. The width ofthe strip line may be between approximately 100 microns andapproximately 1000 microns. In some aspects, the length of the stripline may be between approximately 2 millimeters and approximately 6millimeters. In some cases, the main amplifier and peaking amplifier maybe integrated on a same substrate as the conductive strip line.

In some implementations, the substrate on which the conductive stripline is formed comprises a high-frequency laminate. In some cases, thesubstrate may comprise a semiconductor.

According to some aspects, one or both of the main amplifier and peakingamplifier comprises gallium-nitride transistors.

In some implementations, the impedance inverter consists essentially ofa conductive strip line having a width and a length integrated on asubstrate and bond wires connected between the conductive strip line andoutputs from the main amplifier and the peaking amplifier.

In some aspects, a Doherty amplifier may further comprise animpedance-matching element connected between the combining node and anoutput port of the Doherty amplifier. The impedance-matching element mayprovide an output impedance of 50 ohms. In some implementations, theimpedance-matching element may provide an output impedance betweenapproximately 25 ohms and approximately 100 ohms for the Dohertyamplifier.

In some implementations, a RF fractional bandwidth for the amplifier maybe between approximately 6% and approximately 18%. An operatingfrequency for the Doherty amplifier may be between approximately 500 MHzand approximately 6 GHz. In some cases, a Doherty amplifier may be ratedfor an output power level from the combining node that is betweenapproximately 20 Watts and approximately 100 Watts. In some aspects, aDoherty amplifier may be connected to a cellular transmitter (e.g.,incorporated in apparatus of a cellular phone or cellular base station).

Some embodiments relate to methods for amplifying signals. Embodiedmethods may include acts of splitting a received signal into a firstsignal and a second signal having a first phase with respect to thefirst signal; amplifying the first signal with a main amplifier;amplifying the second signal with a peaking amplifier; providing anoutput from the main amplifier directly to an input of an impedanceinverter, wherein the impedance inverter comprises an integrateddistributed inductor; and introducing a second phase that compensatesfor the first phase with the impedance inverter.

In some aspects, the second phase is not more than 95 degrees.

According to some implementations, a method may further include acts ofcombining an output from the impedance inverter with an output from thepeaking amplifier to produce a combined output; and providing thecombined output to an impedance-matching element. According to someimplementations, a method may further include matching an impedance withthe impedance-matching element to a value between approximately 25 ohmsand approximately 100 ohms. The combining may be done at a drain bondingpad of the peaking amplifier. In some implementations, the integrateddistributed inductor comprises a conductive strip line having a widthand a length integrated on a substrate. The width may be betweenapproximately 100 microns and approximately 1000 microns. The length maybe between approximately 2 millimeters and approximately 6 millimeters.In some cases, the impedance inverter further comprises bond wires orconductive interconnects connected between the conductive strip line andoutputs of the main amplifier and the peaking amplifier.

In some implementations, the main amplifier and peaking amplifier may beintegrated on a same substrate as the conductive strip line. Thesubstrate may comprise a high-frequency laminate. In some aspects, thereceived signal is at a frequency between approximately 500 MHz andapproximately 6 GHz. According to some aspects, an RF fractionalbandwidth for amplifying the signals may be between approximately 6% andapproximately 18%. A power level of the combined output may be betweenapproximately 20 Watts and approximately 100 Watts.

In some aspects, a method may further include providing the combinedoutput for transmission by a cellular base station. A method may alsocomprise providing no impedance-matching elements between the mainamplifier and the impedance inverter that would match an outputimpedance of the main amplifier to 50 ohms. Any of the amplifyingmethods may be performed with gallium-nitride transistors.

Some embodiments relate to a Doherty amplifier comprising an RF input, amain amplifier connected to the RF input, a peaking amplifier connectedto the RF input, a combining node at which an output from the mainamplifier combines with an output from the peaking amplifier, and animpedance inverter comprising a first integrated distributed inductorand a second integrated distributed inductor connected by a capacitor.The impedance inverter may be connected between an output of the mainamplifier and to the combining node.

In some aspects, an input to the impedance inverter comprises one ormore bond wires connected to the output of the main amplifier. Thecombining node may be located at a drain bonding pad of the peakingamplifier. In some implementations, there are no impedance-matchingelements connected between the main amplifier and the impedance inverterto match an output impedance of the main amplifier to 50 ohms. Theimpedance inverter may rotate a phase of a first signal amplified by themain amplifier by no more than 95 degrees with respect to a phase of asecond signal amplified by the peaking amplifier.

According to some implementations, the first distributed inductorcomprises at least one conductive strip line having a first width and afirst length integrated on a substrate. The second distributed inductormay comprise at least one conductive strip line having a second widthand a second length integrated on a substrate. The impedance invertermay further comprise lumped inductive and capacitive elements. In someimplementations, the conductive strip lines may be mainly inductive. Insome cases, the first and second conductive strip lines may haveapproximately equal lengths. In some aspects, the impedance inverter mayfurther comprise bond wires or conductive interconnects connectedbetween the first conductive strip line and an output of the mainamplifier. In some aspects, the impedance inverter may further comprisebond wires or conductive interconnects connected between the secondconductive strip line and an output of the peaking amplifier. The widthof the first and second conductive strip lines may be betweenapproximately 100 microns and approximately 1000 microns. In someaspects, the length of the first or second strip line may be betweenapproximately 1 millimeters and approximately 3 millimeters. In somecases, the main amplifier and peaking amplifier may be integrated on asame substrate as the first and second conductive strip lines.

In some implementations, the substrate on which one or both of theconductive strip lines are formed comprises a high-frequency laminate.In some cases, the substrate may comprise a semiconductor.

According to some aspects, one or both of the main amplifier and peakingamplifier comprises gallium-nitride transistors.

In some implementations, the impedance inverter consists essentially ofa first conductive strip line and a second conductive strip lineconnected by a capacitor, bond wires connected between the firstconductive strip line and an output from the main amplifier, and bondwires connected between the second conductive strip line and an outputfrom the peaking amplifier.

In some aspects, a Doherty amplifier may further comprise animpedance-matching element connected between the combining node and anoutput port of the Doherty amplifier. The impedance-matching element mayprovide an output impedance of 50 ohms. In some implementations, theimpedance-matching element may provide an output impedance betweenapproximately 25 ohms and approximately 100 ohms for the Dohertyamplifier.

In some implementations, a RF fractional bandwidth for the amplifier maybe between approximately 6% and approximately 18%. An operatingfrequency for the Doherty amplifier may be between approximately 500 MHzand approximately 6 GHz. In some cases, a Doherty amplifier may be ratedfor an output power level from the combining node that is betweenapproximately 20 Watts and approximately 100 Watts. In some aspects, aDoherty amplifier may be connected to a cellular transmitter (e.g.,incorporated in apparatus of a cellular phone or cellular base station).

Some embodiments relate to methods for amplifying signals. Embodiedmethods may include acts of splitting a received signal into a firstsignal and a second signal having a first phase with respect to thefirst signal; amplifying the first signal with a main amplifier;amplifying the second signal with a peaking amplifier; providing anoutput from the main amplifier directly to an input of an impedanceinverter, wherein the impedance inverter comprises a first integrateddistributed inductor and a second integrated distributed inductorconnected by a capacitor; and introducing a second phase thatcompensates for the first phase with the impedance inverter.

In some aspects, the second phase is not more than 95 degrees.

According to some implementations, a method may further include acts ofcombining an output from the impedance inverter with an output from thepeaking amplifier to produce a combined output; and providing thecombined output to an impedance-matching element. According to someimplementations, a method may further include matching an impedance withthe impedance-matching element to a value between approximately 25 ohmsand approximately 100 ohms. The combining may be done at a drain bondingpad of the peaking amplifier. In some implementations, the integrateddistributed inductor comprises a first conductive strip line having awidth and a length integrated on a substrate. The width may be betweenapproximately 100 microns and approximately 1000 microns. The length maybe between approximately 1 millimeters and approximately 3 millimeters.In some cases, the impedance inverter further comprises bond wires orconductive interconnects connected between the first conductive stripline and an output of the main amplifier.

In some implementations, the main amplifier and peaking amplifier may beintegrated on a same substrate as the first conductive strip line. Thesubstrate may comprise a high-frequency laminate. In some aspects, thereceived signal is at a frequency between approximately 500 MHz andapproximately 6 GHz. According to some aspects, an RF fractionalbandwidth for amplifying the signals may be between approximately 6% andapproximately 18%. A power level of the combined output may be betweenapproximately 20 Watts and approximately 100 Watts.

In some aspects, a method may further include providing the combinedoutput for transmission by a cellular base station. A method may alsocomprise providing no impedance-matching elements between the mainamplifier and the impedance inverter that would match an outputimpedance of the main amplifier to 50 ohms. Any of the amplifyingmethods may be performed with gallium-nitride transistors.

The foregoing apparatus and method embodiments may be implemented withany suitable combination of aspects, features, and acts described aboveor in further detail below. These and other aspects, embodiments, andfeatures of the present teachings can be more fully understood from thefollowing description in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The skilled artisan will understand that the figures, described herein,are for illustration purposes only. It is to be understood that in someinstances various aspects of the embodiments may be shown exaggerated orenlarged to facilitate an understanding of the embodiments. The drawingsare not necessarily to scale, emphasis instead being placed uponillustrating the principles of the teachings. In the drawings, likereference characters generally refer to like features, functionallysimilar and/or structurally similar elements throughout the variousfigures. Where the drawings relate to microfabricated circuits, only onedevice and/or circuit may be shown to simplify the drawings. Inpractice, a large number of devices or circuits may be fabricated inparallel across a large area of a substrate or entire substrate.Additionally, a depicted device or circuit may be integrated within alarger circuit.

When referring to the drawings in the following detailed description,spatial references “top,” “bottom,” “upper,” “lower,” “vertical,”“horizontal,” and the like may be used. Such references are used forteaching purposes, and are not intended as absolute references forembodied devices. An embodied device may be oriented spatially in anysuitable manner that may be different from the orientations shown in thedrawings. The drawings are not intended to limit the scope of thepresent teachings in any way.

FIG. 1 depicts a first arrangement of a Doherty amplifier;

FIG. 2 depicts an equivalent circuit for a Doherty amplifier;

FIG. 3 shows different frequency-response curves and RF fractionalbandwidths for different Doherty amplifier designs, according to someembodiments;

FIG. 4 depicts a modified Doherty amplifier in which signals from themain and peaking amplifiers are combined before impedance is matched tothe load, according to some embodiments;

FIG. 5A depicts elements of an impedance inverter that includes lumpedand integrated distributed inductance, according to some embodiments;

FIG. 5B depicts elements of a power amplifier, according to someembodiments;

FIG. 5C depicts elements of an impedance inverter that includes lumpedand integrated distributed inductance and lumped capacitance, accordingto some embodiments;

FIG. 6 indicates bandwidth characteristics for different Dohertyamplifier designs;

FIG. 7 depicts an impedance-matching element at an output of a modifiedDoherty amplifier, according to some embodiments; and

FIG. 8 depicts a double-section impedance-matching element at an outputof a modified Doherty amplifier, according to some embodiments.

Features and advantages of the illustrated embodiments will become moreapparent from the detailed description set forth below when taken inconjunction with the drawings.

DETAILED DESCRIPTION

As described above, amplifiers comprising gallium nitride (GaN)transistors are useful for high-speed, high-voltage, and high-powerapplications because of the favorable material properties of galliumnitride. In some cases, transistors formed from other semiconductormaterials such as gallium arsenide, silicon carbide, silicon germanium,etc., may be suitable for certain high-speed, high-voltage, andhigh-power applications. Technology areas in which GaN transistors arefinding increasing use are radio-frequency (RF) communications andradar. In RF communications, for example, GaN transistors may be used inDoherty amplifiers at a base station to amplify data signals forwireless broadcasting within a cell covered by the base station.

One arrangement of a Doherty amplifier 100 is shown in FIG. 1 . ADoherty amplifier may comprise a main amplifier 132 and a peakingamplifier 138 arranged on parallel circuit branches. An input RF signalis split by a 90-degree coupler 110 that provides an in-phase attenuatedsignal to the main amplifier and an attenuated signal rotated by 90degrees (typically delayed by 90°) to the peaking amplifier. Afteramplification, an impedance inverter 150 that includes a compensating90-degree rotation is used to recombine the two signals into a combinedand amplified output RF signal. An output impedance-matching element 160may be connected to the combining node to match the output impedance ofthe Doherty amplifier to the impedance of a load (not shown).

Impedance-matching components 122, 124 may be placed before the mainamplifier and peaking amplifier in a Doherty amplifier. These matchingcomponents may be used to match the impedances of the transmission linesfrom the 90-degree coupler 110 to the input impedances of the twoamplifiers, so that signal reflections from the amplifiers are reducedor essentially eliminated. Additional impedance-matching components 142,144 may be placed at the outputs of the main and peaking amplifiers tomatch impedances to the input of the impedance inverter 150 (which maybe 50 ohms by design) and to the combining node 155. Theimpedance-matching components 142, 144 may comprise resistive,capacitive, and/or inductive circuit elements.

The inventor has recognized and appreciated that there is a cost inbandwidth performance of a Doherty amplifier 100 when impedance-matchingcomponents 142, 144 are placed between the outputs of the main andpeaking amplifiers 132, 138 and the impedance inverter 150 and combiningnode 155. At these locations, the impedance-matching components 142, 144add electrical path length between the two amplifiers so that it may notbe possible for the impedance inverter 150 to employ only a 90-degreerotation to compensate for the phase rotation introduced by the 90°coupler. Instead, the impedance inverter 150 may need to operate with aphase rotation θ that is an odd integer multiple of 90 degrees accordingto the following relationθ˜(2n+1)90°  EQ. 1where n is an integer value of 1 or greater.

To investigate the cost in bandwidth performance of a Doherty amplifier100 due to the impedance-matching elements 142, 144, high-frequencysimulations were performed using a low-power circuit model 200, which isdepicted in FIG. 2 . The low-power circuit model represents a case whenthe peaking amplifier is off. The inventor has recognized andappreciated that when the peaking amplifier is off, a substantialimpedance mismatch can occur between the output of the main amplifier132 and the combining node 155 in the Doherty amplifier. Accordingly,the low-power operation may constrain the rated RF fractional bandwidthfor a Doherty amplifier, e.g., a guaranteed bandwidth for all signallevels. In the low-power circuit model 200, the main amplifier 132 isrepresented as a first current source I_(m) and the peaking amplifier138 is represented as a second current source I_(p), which outputs nocurrent. The impedance inverter 150 is modeled as a transmission linehaving a resistance R_(o) and having an adjustable phase rotation, whichcan be set to an odd multiple of 90° at the center frequency ofoperation (2 GHz for this simulation). The load impedance is R_(o)/2.For purposes of the simulation, the impedance of the peaking amplifierwhen off is given a value of 20R_(o).

Simulations circuits and circuit elements described herein may beimplemented using a software tool such as Advanced Design System (ADS)available from Keysight Technologies, Inc. of Santa Rosa, Calif. Othersuitable software tools include, but are not limited to NI AWR DesignEnvironment available from AWR Corporation of El Segundo, Calif. andSonnet® software tools available from Sonnet Software of North Syracuse,N.Y.

Results from the simulations of the Doherty amplifier are shown in FIG.3 . The frequency-response curves 310, 320, 330 plotted in the graphrepresent the scattering parameter S(1,1) evaluated looking from theoutput of the main amplifier 132 (e.g. the current source I_(m)) intothe impedance inverter 150. The frequency-response curves represent anamount of signal reflected back to the main amplifier (e.g.,voltage-to-standing-wave ratio) as a function of frequency. For purposesof evaluating amplifier performance, an RF fractional bandwidth(Δω/ω_(o)) for the amplifier may be determined from a frequencydifference Δω between the −20 dB points on the frequency-response curveswhere the value of the back-reflected signal is at least 20 dB below thesignal level input to the impedance inverter.

When the impedance-matching elements 142, 144 are located before theimpedance inverter and combining node, the minimum allowable phaserotation by the impedance inverter 150 may be 270° due to the extraelectrical path length added by the impedance-matching elements. Onesuch case (n=1 in EQ. 1, dashed line) corresponds to thefrequency-response curve 320 plotted in FIG. 3 . In this case, the RFfractional bandwidth is approximately 6%. If the added electrical pathintroduced by the impedance-matching elements 142, 144 is greater, theminimum allowable phase introduced by the impedance inverter 150 mayincrease to 450° (n=2), which results in the frequency-response curve310. For this case, the RF fractional bandwidth reduces to about 3%.Conventional Doherty amplifiers for RF communication systems typicallyoperate with RF fractional bandwidths less than about 4%. On the otherhand, if the minimum allowable phase introduced by the impedanceinverter 150 were 90°, then the RF fractional bandwidth could increaseto over 17% as indicated by the frequency-response curve 330.

The inventor has recognized and appreciated that removing theimpedance-matching elements 142, 144 before the impedance inverter 150and combining node 155 allows a reduction in the compensating phaseintroduced by the impedance inverter to 90° or approximately 90°.Although the compensating phase angle is preferably 90°, in some casesthe coupler 110 may impart a phase difference between 85° and 95°, whichis compensated by the impedance inverter.

FIG. 4 depicts an embodiment of a modified Doherty amplifier 400 inwhich signals from the main and peaking amplifiers are combined first,and then impedance is matched to a load after combining. For example,impedance matching can be accomplished in an output impedance-matchingelement 420 located after the combining node 155. According to someembodiments, the combining node 155 may be located at the output of thepeaking amplifier 138. An input to an impedance inverter 410 may connectdirectly to an output from the main amplifier 132. There may be noimpedance-matching element between the output from the main amplifierand an input to the impedance inverter 410 that matches or rotates theimpedance from the main amplifier to 50 ohms, for example. Further,there may be no impedance-matching element between the output of thepeaking amplifier 138 and the combining node 155.

Further details of an impedance inverter 410 and modified Dohertyamplifier 400 are depicted in FIG. 5A, according to some embodiments. Insome cases, the impedance inverter 410 comprises a conductive strip line510 (e.g., a microstrip line) that extends a length L. The length L mayextend between and along output drain bonding pads 533 of the mainamplifier 132 and the peaking amplifier 138. The conductive strip line510 may have a width W. The length of the conductive strip line may bebetween approximately 2 millimeters and approximately 6 millimeters,according to some embodiments, and may be selected to provide a desiredinductance for the strip line 510. The width of the conductive stripline may be between approximately 100 microns and approximately 1000microns, according to some embodiments, and may be selected to provide adesired inductance for the strip line. In some implementations, theconductive strip line is formed over a ground conductor or ground planeand separated from the ground conductor or ground plane by a dielectricmaterial (not shown). In other embodiments, the conductive strip linemay not be formed over or adjacent to a ground plane. Instead, a groundplane may be removed from an area of a PCB at which the conductive stripline is patterned. The conductive strip line, when implemented in theimpedance inverter for RF signals, may comprise an integrateddistributed impedance element which is essentially entirely inductive.In some implementations, the strip line may include some parasiticcapacitance and resistance.

The conductive strip line may be formed on a substrate 505, upon whichan output impedance matching element 560 may be manufactured. In someembodiments, the main amplifier 132 and the peaking amplifier 138 may bemounted adjacent to the substrate 505 and be on one or more separatedies. In some implementations, the conductive strip line 510 may beintegrated onto a same substrate on which the main amplifier 132 and/orthe peaking amplifier 138 are formed. The substrate 505 on which theconductive strip line is formed may comprise a printed circuit board insome embodiments, a high-frequency laminate capable of carrying signalsat GHz frequencies in some embodiments, a ceramic, or a semiconductor.An example of a high-frequency laminate is laminate model RO4003®available from Rogers Corporation of Chandler, Ariz.

According to some embodiments, an impedance inverter 410 may furtherinclude one or more amplifier output bond wires 520 that connect to adrain bond pad 533 of the main amplifier and the conductive strip line510 near a first end of the strip line (e.g., located within about afirst ⅓ of the length of the strip line). Additionally, there may be oneor more amplifier output bond wires 520 connected between a drain bondpad of the peaking amplifier 138 and an opposing end of the conductivestrip line 510. The output bond wires 520 may be arranged at essentiallyuniform spacing along the strip line in some embodiments, but may bearranged non-uniformly in other embodiments. The spacing between thebond wires may be between approximately 100 microns and approximately800 microns. The bond wires 520 may be comprise gold or any othersuitable conductor, may have a diameter between 20 microns and 80microns, and may arc or extend over the substrate 505 and substrate 503to a height between approximately 50 microns and approximately 250microns. The output bond wires 520 comprise lumped inductive elements ofthe impedance inverter 410. Such bond wires are recognized in the fieldof RF electronics as “lumped inductors” having an inductance that isdetermined primarily by a length and diameter of the bond wire. Theremay be amplifier input bond wires 540 connecting to gate bond pads 531of the main amplifier 132 and the peaking amplifier 138.

In some embodiments where the conductive strip line 510, main amplifierand/or peaking amplifier are integrated onto a same substrate, bondwires 520 may not be used. Instead, conductive interconnects such amicrostrip transmission lines or conductive traces may be used toconnect the strip line 510 to outputs from the main and peakingamplifiers. In some implementations where the conductive strip line 510,main amplifier and/or peaking amplifier are integrated onto a samesubstrate, one or both drain bond pads 530 may be replaced with orsubsumed into the conductive strip line 510, so that the inductance ofthe impedance inverter is essentially entirely a distributed inductance.

For the embodiment depicted in FIG. 5A, a combining node of the Dohertyamplifier 400 may be located at the drain bond pad 533 of the peakingamplifier 138. In such embodiments, the impedance inverter 410 maycomprise lumped inductive elements (for example, the main and peakingamplifier output bond wires 520) and an integrated distributed inductiveelement comprising the conductive strip line 510. For purposes ofanalyzing RF performance, the impedance inverter may include lumpedcapacitive elements, which may include the drain-to source capacitancesof the main amplifier 132 and the peaking amplifier 138 and capacitanceof the drain bond pads 533. The impedance inverter 410 may furtherinclude a small distributed capacitance of the conductive strip line510.

In some implementations, lumped capacitance elements may be added asshunts to the drain bond pads 533 and/or inductive strip line 510 toadjust an operating frequency of the Doherty amplifier to a desiredvalue, or added in series to extend a length of the impedance inverterfor higher power applications. In some cases, an integrated, inductivestrip line may comprise two separated strip lines 512 that are connectedby a capacitor 580 (e.g., a surface mount capacitor) added in seriesbetween the two halves of the strip line, as depicted in FIG. 5C. Thisarrangement of two strip lines can extend the overall distance betweenthe two amplifiers, allowing larger amplifiers 132, 138 and higher powercapability, without adding more inductance. However, the addedcapacitance should be limited to avoid altering phase rotation in theimpedance inverter beyond 95 degrees.

In some cases, there may be output bond wires 550 connected between adrain bond pad 533 of the peaking amplifier 138 and an outputimpedance-matching element 560 of the Doherty amplifier. The outputimpedance-matching element 560 may comprise lumped and/or distributedimpedance elements that are used to match an impedance from the drainbond pad 533 of the peaking amplifier 138 to a load impedance (e.g., 50ohms) at a load plane 570.

Additional details of structure near the drain bond pads 533 of the mainor peaking amplifier are shown in FIG. 5B, for some embodiments. Themain amplifier 132 and/or the peaking amplifier 138 may comprise alinear array of transistors having gate conductors 532, drain contacts534, and source contacts 536 formed on a semiconductor substrate 503.The drain contacts 534 for an amplifier may connect to a drain bond pad533, at which one or more output bond wires 520, 550 may be bonded. Insome implementations, the active regions of the transistors may comprisegallium nitride, which is desirable for high-power, high-frequencyamplification of RF signals as described above. As used herein, thephrase “gallium nitride” refers to gallium nitride (GaN) and any of itsalloys, such as aluminum gallium nitride (Al_(x)Ga_((1-x))N), indiumgallium nitride (In_(y)Ga_((1-y))N), aluminum indium gallium nitride(Al_(x)In_(y)Ga_((1-x-y))N), gallium arsenide phosporide nitride(GaAs_(x)P_(y)N_((1-x-y))), aluminum indium gallium arsenide phosporidenitride (Al_(x)In_(y)Ga_((1-x-y))As_(a)P_(b)N_((1-a-b))), amongstothers. In some cases, the transistors may be formed from othersemiconductor materials such as gallium arsenide, silicon carbide,silicon germanium, silicon, indium phosphide, etc. and the invention isnot limited to gallium-nitride-based amplifiers.

A benefit of a conductive strip line 510 in the impedance inverter as aninductive impedance element is that it can more readily allow forscalability of power of the Doherty amplifier 400 compared to lumpedinductive elements only. For example, the power-handling capability of aDoherty amplifier may be determined by the size of transistors in themain amplifier 132 and peaking amplifier 138. Power may be increased ina Doherty amplifier by increasing the number of transistors (gateconductors, drain contacts, and source contacts) along the linear arrayof transistors in the main amplifier and the peaking amplifier. However,increasing the number of transistors and length of the arrays canrequire additional amplifier output bond wires 520 between the twoamplifiers and corresponding locations on the conductive strip line 510,and may require increasing the length of the strip line.

The addition of amplifier output bond wires 520 and increased length ofthe strip line would normally increase the inductance of the impedanceinverter 410. The inventor has recognized and appreciated that thisincrease in inductance may be offset by decreasing the inductance of theconductive strip line 510. Inductance of the strip line 510 may bedecreased by increasing its width W. By selecting the length and widthof the strip line, the distributed inductance of the strip line 510 maybe tuned to a desired value. According to some embodiments, a total ofthe distributed inductance of the strip line may be betweenapproximately 250 picoHenries and approximately 1.5 nanoHenries.

For power scaling in some cases, the inductance of the strip line 510may be decreased by increasing its width W and/or decreasing its lengthL. Conversely, the inductance of the strip line may be increased bydecreasing its width W and/or increasing its length L. Such changes willalso affect any capacitance and resistance of the strip line. Theconductive strip line 510 comprises a tunable impedance element for theimpedance inverter 410 that may be adjusted at the patterning stage ofmanufacture for a desired application. Accordingly, power of the Dohertyamplifier 400 may be scaled while preserving an operating frequency andbandwidth performance of the Doherty amplifier 400. Such scalabilitywould not be possible in a purely lumped-element impedance inverterwhere the drain bond pad 533 of the main amplifier 132 is wire bondeddirectly to the drain bond pad of the peaking amplifier 138.

Adding length to the transistor arrays may also add electrical pathlength to the impedance inverter 410. As a result, there will be a limitto the total allowed electrical path length, and consequently power,that the Doherty amplifier 400 can handle when arranged as depicted inFIG. 5A. Essentially, the electrical path length can be increased untilthe phase rotation reaches approximately 90 degrees, though highervalues (e.g., up to 95 degrees) may be possible in some cases where thecoupler 110 provides a higher phase rotation than 90 degrees. Becausethe phase rotation for a physical path length will depend on frequency,lower-frequency devices may allow greater length extensions of theamplifier transistor arrays and therefore handle high powers. Initialcalculations indicate that Doherty amplifiers configured as shown inFIG. 5A should be capable of amplifying RF signals in frequency rangesbetween about 500 MHz and about 6 GHz to power levels between about 5Watts and about 100 Watts at 500 MHz and between about 5 Watts and about35 Watts at 6 GHz. In some implementations, the rated output powerlevels can be as high as between about 20 Watts and about 100 Watts at500 MHz and between about 20 Watts and about 35 Watts at 6 GHz.

In an alternative embodiment, the power capability of the Dohertyamplifier 400 may be doubled. Referring again to FIG. 5A, a second mainamplifier 132 may be located on a side of the conductive strip lineopposite the illustrated first main amplifier 132. The outputimpedance-matching element 560 may be rotated 90 degrees and mountednear the end of the conductive strip line 510 by the peaking amplifier138. A second peaking amplifier 138 may be located on a side of theconductive strip line opposite the illustrated first peaking amplifier138. Drain bond pads from the additional main and peaking amplifiers maybe wire bonded to the conductive strip line. Additional bond wires maybe connected at angles from the output impedance-matching element 560 todrain bond pads of the peaking amplifiers 138.

Several circuit simulations were carried out for a Doherty amplifier 400as arranged in FIG. 4 , of which some results are shown in FIG. 6 . In afirst simulation, the impedance inverter 410 was modeled using alumped-equivalent model: a single lumped inductor and shunt capacitorsarranged in a pi network connected between the main amplifier 132 andthe peaking amplifier 138. The capacitors were connected as shunts oneither side of the inductor. The value of the inductor was 1.04 nH. Thevalues of the two capacitors were 1.99 pF, which represented a sum ofthe drain-to-source capacitances (˜1.6 pF) and drain bond padcapacitance (˜0.39 pF). The circuit arrangement was similar to thatshown in FIG. 2 , except the impedance inverter 150 is replaced with thelumped pi network and the peaking current source I_(p) is replaced witha resistance of 20R_(o). The value of R_(o) was 22.9 ohms. This firstsimulation was carried out to analyze the feasibility of the Dohertyamplifier 400 in which combining is carried out first before impedancematching.

In some embodiments, the values of a Doherty amplifier's operatingfrequency φ_(o) and inductance Ls of the strip line 510 are constrainedin part by amplifier design. For example, an amplifier design may have adrain-to-source capacitance C_(ds), and be rated at a maximumdrain-to-source current I_(max) for an operating voltage V_(ds). Theresistance R_(o) at which maximum power may be transferred from theamplifier may be determined approximately from the following relation.R _(o)≈2(V _(ds) −V _(k))/I _(max)  (EQ. 2)where V_(k) is the knee voltage for the amplifier. Once R_(o) isestimated, then it is desirable to have the admittance of the shuntcapacitance C_(sh) (primarily determined by C_(ds), though it mayinclude drain pad capacitance and any added capacitance) and theimpedance of the impedance inverter's inductance L_(c) (determined fromthe wire bonds 520 and strip line 510) match the correspondingadmittance and impedance values of R_(o), which yields:R _(o)≈1/C _(sh)ω_(o)  (EQ. 3)R _(o)≈ω_(o) L _(c)  (EQ. 4)Since C_(ds) is primarily determined by the amplifiers' design and maybe the dominant capacitance, EQ. 3 roughly constrains the operatingfrequency of the amplifier, though it may be tuned downward by addingadditional shunt capacitance. According to some embodiments, when theoperating frequency is selected, the conductive strip line may bedesigned to provide inductance according to EQ. 4.

A frequency-response curve 610 (dotted curve) from the first simulationwith a lumped-element impedance inverter is plotted in FIG. 6 . The plotrepresents the scattering parameter S(1,1) looking into the impedanceinverter (e.g., looking into the pi network at the first capacitiveshunt). The response shows a bandwidth of about 400 MHz centered at anoperating frequency of approximately 3.5 GHz. This bandwidth is greaterthan 11% and represents a significant improvement over a comparablebandwidth performance of a conventional Doherty amplifier at RFfrequencies, which is typically less than about 4%.

In a second simulation, the lumped inductor was replaced with adistributed inductor that more accurately modeled the integratedconductive strip line 510 depicted in FIG. 5A. For this simulation,modeling of electromagnetic waves at different frequencies carried bythe conductive strip line 510 was carried out using an electromagnetic(EM) field simulation tool. In the EM simulation, the conductive stripline was modeled as having six input ports corresponding to the bondwires 520. The input ports were 50 μm wide, with three spaced at eachend on a 500 μm pitch. The length of the conductive strip line was 3.7mm and the width was 300 μm. The conductive strip line was modeled asbeing formed from copper (17.5 μm thick) on a high-frequency laminatehaving a dielectric constant of 3.55 and a loss tangent of 0.002. Thethickness of the laminate separating the conductive strip line from aground plane was 305 μm. For the EM simulation, a mesh was used having50 cells per wavelength at 4 GHz. The results from the EM simulation forthe strip line 510 were used in a circuit simulation of the impedanceinverter in which the same values of lumped capacitances (1.99 pF) wereused and arranged in the pi network. The circuit arrangement wasotherwise the same as that used to generate the frequency-response curve610. Results from this second simulation are plotted as thefrequency-response curve 620 (dashed curve), which indicates that usinga distributed inductive element in the impedance inverter adds minimalreduction in the RF fractional bandwidth compared to a purely lumpedelement impedance inverter. Accordingly, the inductive strip line 510enables power scalability while essentially maintaining operatingfrequency and RF fractional bandwidth performance.

Additional EM simulations were carried out to more accurately representthe output bonding pads 533 of the main amplifier 132 and peakingamplifier 138 as well as a circuit simulation to represent the bondwires 520 connected to the bonding pads 533. For the EM simulations, thebonding pads 533 measured approximately 1.8 mm by approximately 85microns. The bond wires were represented as having a 25 micron diameter,a conductivity of 5×10⁷ Siemens, extending over a gap of about 500microns and rising to a maximal height of about 150 microns above theamplifier die. Using the results from the EM simulations in the circuitmodel for the Doherty amplifier 400 did not appreciably alter thefrequency response curve 620.

In an actual device, impedance at the output of the impedance inverter410 may need to be matched to impedance of a load (e.g., 50 ohms). Tofurther evaluate the performance of the Doherty amplifier 400, an outputimpedance-matching element 560 was added to the circuit and simulationscarried out to account for the added element. For these simulations, anoutput impedance-matching element 560 depicted in FIG. 7 was used, butthe depicted element is only one example of an output-impedance matchingelement and the invention is not limited to only this configuration.Other embodiments may be used for the output-impedance-matching elementin other implementations.

According to some embodiments, output bond wires 550 may be bonded to anoutput strip line 710 of the output impedance-matching element 560.Shunt capacitors 712, 714 may connect between the output strip line 710and pads 720, which are connected to an underlying ground conductorusing a via and shunt conductor 730. An output capacitor 718 may connectbetween the output strip line 710 and an output bonding pad 750. For thesimulation, the output bonding pad 750 may be shunted to ground with a50 ohm resistive via to emulate a load. The length and width of theoutput strip line 710, the values of the shunt capacitors 712, 714, andthe value of the output capacitor 718 may be selected to match animpedance from the combining node to an impedance at the load plane 570.

Results of a simulation of amplifier performance that includes an outputimpedance-matching element 560 as arranged in FIG. 7 is plotted in FIG.6 as the frequency-response curve 630. For this simulation, an impedanceat the combining node (approximately 11.45 ohms) was matched to a loadimpedance of approximately 50 ohms. In an EM simulation, the strip line710 measured approximately 1.4 mm in length with a width ofapproximately 350 microns, and otherwise used the same electromagneticproperties that were used for the conductive strip line 510. The shuntcapacitors 712, 714 were modeled as surface-mount devices (SMDs) havinga capacitance of 0.75 pF each. The shunt capacitors and conductive vias730 had a combined resistance of 0.15 ohm and inductance of 0.3 nH each.The output capacitor 718 was also modeled as an SMD having a capacitanceof 6.8 pF, which had a combined resistance of 0.15 ohm and inductance of0.3 nH.

The result of the simulation that includes the effect of the outputimpedance-matching element 560, and also includes EM simulations of theoutput bonding pads 533, is plotted in the frequency-response curve 630of FIG. 6 . For the illustrated impedance-matching element, the RFfractional bandwidth of the amplifier reduces to approximately 200 MHzor about 6% at an operating frequency of about 3.5 GHz. Even with thisreduction, the RF fractional bandwidth for the modified Dohertyamplifier is nearly twice the bandwidth of a conventional Dohertyamplifier. The results of this simulation indicate that if the outputimpedance-matching is not done well, or has a narrow RF fractionalbandwidth, then the overall bandwidth of the device may be limited bythe output impedance-matching element 560.

To recover a broader bandwidth, a double-section outputimpedance-matching element 800 may be used, as depicted in FIG. 8 . Adouble-section impedance-matching element may comprise an addedinductive strip line 850 that connects to the output bonding pad 750 andto capacitive shunt 814. The dimensions of the strip line 710 may beresized to provide a desired inductance for the first section.

Some embodiments may include transistor biasing components comprising aninductive strip line 840 that connects to a DC biasing port 830, atwhich voltage for biasing drains of transistors in the amplifiers 132,138 may be applied. A shunt capacitor 816 may be connected to thebiasing port 830. When installed in a device, an additional capacitormay be mounted external to the board on which the impedance-matchingelement 800 is formed and arranged in parallel to the shunt capacitor816. The external capacitor may have a value between 2 microFarads and50 microFarads.

Further simulations were carried out for the double-sectionimpedance-matching element in which the values of capacitances were asfollows: C1=C2=0.7 pF, C3=1.2 pF, and C4=6.8 pF. The double-sectionimpedance-matching element 800 provides improved impedance matching overa range of RF frequencies near the center or carrier frequency at 3.5GHz, as compared to the single section depicted in FIG. 7 . Therefore,it removes a bandwidth bottleneck associated with the single sectionimpedance-matching element 560 and recovers the RF fractional bandwidthavailable for the impedance inverter. The simulations show that theresulting RF fractional bandwidth recovers to approximately 18%.

In some implementations, additional impedance-matching sections may beincluded between the impedance inverter and load. Whether comprising oneor more sections, an output impedance-matching element preferablytransforms the impedance at the output of the impedance inverter 410 tomatch or approximately match the impedance at the load plane 570 over abandwidth of interest (e.g., 80 MHz, 100 MHz, 200 Mhz, 400 MHz, or anydesired RF fractional bandwidth in this range) at the carrier frequency(3.5 GHz in the above example, though other carrier frequencies may beused).

Methods for operating a Doherty amplifier using the above-describedapparatus are also contemplated. In some implementations, a method foroperating a Doherty amplifier 400 may comprise acts of splitting areceived signal into a first signal and a second signal having a firstphase with respect to the first signal, amplifying the first signal witha main amplifier 132, and amplifying the second signal with a peakingamplifier 138. A method embodiment may further comprise providing anoutput from the main amplifier directly to an input of an impedanceinverter 410, wherein the impedance inverter comprises an integrateddistributed inductor, and introducing a second phase with the impedanceinverter that compensates for the first phase. In some implementations,a method for operating a Doherty amplifier may further comprisecombining an output from the impedance inverter 410 with an output fromthe peaking amplifier 138 to produce a combined output, and providingthe combined output to an impedance-matching element 560 that matchesthe output impedance to the impedance of a load. The load impedance mayhave a value of 50 ohms or approximately 50 ohms. In someimplementations, the load impedance may have a value betweenapproximately 25 ohms and approximately 100 ohms. Operation of a Dohertyamplifier 400 may further comprise providing the combined output fortransmission by a cellular base station.

CONCLUSION

The terms “approximately” and “about” may be used to mean within ±20% ofa target dimension in some embodiments, within ±10% of a targetdimension in some embodiments, within ±5% of a target dimension in someembodiments, and yet within ±2% of a target dimension in someembodiments. The terms “approximately” and “about” may include thetarget dimension.

The technology described herein may be embodied as a method, of which atleast some acts have been described. The acts performed as part of themethod may be ordered in any suitable way. Accordingly, embodiments maybe constructed in which acts are performed in an order different thandescribed, which may include performing some acts simultaneously, eventhough described as sequential acts in illustrative embodiments.Additionally, a method may include more acts than those described, insome embodiments, and fewer acts than those described in otherembodiments.

Having thus described at least one illustrative embodiment of theinvention, various alterations, modifications, and improvements willreadily occur to those skilled in the art. Such alterations,modifications, and improvements are intended to be within the spirit andscope of the invention. Accordingly, the foregoing description is by wayof example only and is not intended as limiting. The invention islimited only as defined in the following claims and the equivalentsthereto.

What is claimed is:
 1. A Doherty amplifier comprising: an RF input; amain amplifier connected to the RF input; a peaking amplifier connectedto the RF input; an impedance inverter comprising an integrateddistributed inductor connected between an output of the main amplifierand an output of the peaking amplifier; and an output impedance-matchingelement comprising an output strip line, a shunt capacitor connectedbetween the output strip line and ground, an output capacitor connectedbetween the output strip line and an output bonding pad, and aninductive strip line connected between the output bonding pad andground.
 2. The Doherty amplifier of claim 1, wherein: the shuntcapacitor is connected between the output strip line and a via toground; and the inductive strip line is connected between the outputbonding pad and the via to ground.
 3. The Doherty amplifier of claim 1,wherein the impedance inverter comprises one or more bond wiresconnected to the output of the main amplifier.
 4. The Doherty amplifierof claim 3, wherein the impedance inverter further comprises one or moreadditional bond wires connected to the output the peaking amplifier. 5.The Doherty amplifier of claim 1, wherein there are noimpedance-matching elements connected between the main amplifier and theimpedance inverter.
 6. The Doherty amplifier of claim 1, wherein theimpedance inverter rotates a phase of a first signal amplified by themain amplifier by no more than 95 degrees with respect to a phase of asecond signal amplified by the peaking amplifier.
 7. The Dohertyamplifier of claim 1, wherein the integrated distributed inductorcomprises a conductive strip line on a substrate.
 8. The Dohertyamplifier of claim 7, wherein a width of the conductive strip line isbetween 100 and 1000 microns.
 9. The Doherty amplifier of claim 8,wherein a length of the conductive strip line is between 2 and 6millimeters.
 10. The Doherty amplifier of claim 1, wherein theintegrated distributed inductor comprises a first integrated distributedinductor, a second integrated distributed inductor, and a capacitorconnected in series between the first integrated distributed inductorand the second integrated distributed inductor.
 11. The Dohertyamplifier of claim 1, further comprising: a combining node at which theoutput of the main amplifier combines with the output of the peakingamplifier, wherein the combining node comprises a drain bonding pad ofthe peaking amplifier.
 12. The Doherty amplifier of claim 1, wherein oneor both of the main amplifier and peaking amplifier comprisesgallium-nitride transistors.
 13. The Doherty amplifier of claim 1,wherein the impedance inverter consists essentially of: a conductivestrip line having a width and a length integrated on a substrate; bondwires connected between the conductive strip line and outputs from themain amplifier and the peaking amplifier; and drain-to-sourcecapacitances of the main amplifier and the peaking amplifier.
 14. TheDoherty amplifier of claim 1, wherein the output impedance-matchingelement provides an output impedance of approximately 50 ohms for theDoherty amplifier.
 15. The Doherty amplifier of claim 1, wherein an RFfractional bandwidth for the Doherty amplifier is between approximately6% and approximately 18%.
 16. The Doherty amplifier of claim 1, whereinan operating frequency for the Doherty amplifier is betweenapproximately 500 MHz and approximately 6 GHz.
 17. The Doherty amplifierof claim 1, wherein a rated output power level of the amplifier isbetween 20 and 100 Watts.
 18. A method for amplifying signalscomprising: splitting a received signal into a first signal and a secondsignal having a first phase with respect to the first signal; amplifyingthe first signal with a main amplifier; amplifying the second signalwith a peaking amplifier; providing an output from the main amplifier toan input of an impedance inverter, wherein the impedance invertercomprises an integrated distributed inductor; introducing a second phasethat compensates for the first phase with the impedance inverter;combining an output from the impedance inverter with an output from thepeaking amplifier; and providing the combined output to an outputimpedance-matching element, the output impedance-matching elementcomprising an output strip line, a shunt capacitor connected between theoutput strip line and ground, an output capacitor connected between theoutput strip line and an output bonding pad, and an inductive strip lineconnected between the output bonding pad and ground.
 19. The method ofclaim 18, wherein: the shunt capacitor is connected between the outputstrip line and a via to ground; and the inductive strip line isconnected between the output bonding pad and the via to ground.